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| Micropower, Single- and Dual-Supply, Rail-to-Rail Instrumentation Amplifier AD627 FEATURES Micropower, 85 A maximum supply current Wide power supply range (+2.2 V to 18 V) Easy to use Gain set with one external resistor Gain range 5 (no resistor) to 1000 Higher performance than discrete designs Rail-to-rail output swing High accuracy dc performance 0.03% typical gain accuracy (G = +5) (AD627A) 10 ppm/C typical gain drift (G = +5) 125 V maximum input offset voltage (AD627B dual supply) 200 V maximum input offset voltage (AD627A dual supply) 1 V/C maximum input offset voltage drift (AD627B) 3 V/C maximum input offset voltage drift (AD627A) 10 nA maximum input bias current Noise: 38 nV/Hz RTI noise @ 1 kHz (G = +100) Excellent ac specifications AD627A: 77 dB minimum CMRR (G = +5) AD627B: 83 dB minimum CMRR (G = +5) 80 kHz bandwidth (G = +5) 135 s settling time to 0.01% (G = +5, 5 V step) FUNCTIONAL BLOCK DIAGRAM RG -IN +IN -VS 1 2 3 4 AD627 8 7 6 5 RG +VS OUTPUT REF 00782-001 Figure 1. 8-Lead PDIP (N) and SOIC_N (R) 100 90 80 70 CMRR (dB) AD627 60 50 40 30 20 10 1 10 100 FREQUENCY (Hz) 1k 10k 00782-002 TRADITIONAL LOW POWER DISCRETE DESIGN 0 APPLICATIONS 4 to 20 mA loop-powered applications Low power medical instrumentation--ECG, EEG Transducer interfacing Thermocouple amplifiers Industrial process controls Low power data acquisition Portable battery-powered instruments Figure 2. CMRR vs. Frequency, 5 VS, Gain = +5 GENERAL DESCRIPTION The AD627 is an integrated, micropower instrumentation amplifier that delivers rail-to-rail output swing on single and dual (+2.2 V to 18 V) supplies. The AD627 provides excellent ac and dc specifications while operating at only 85 A maximum. The AD627 offers superior flexibility by allowing the user to set the gain of the device with a single external resistor while conforming to the 8-lead industry-standard pinout configuration. With no external resistor, the AD627 is configured for a gain of 5. With an external resistor, it can be set to a gain of up to 1000. A wide supply voltage range (+2.2 V to 18 V) and micropower current consumption make the AD627 a perfect fit for a wide range of applications. Single-supply operation, low power consumption, and rail-to-rail output swing make the AD627 Rev. D Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. ideal for battery-powered applications. Its rail-to-rail output stage maximizes dynamic range when operating from low supply voltages. Dual-supply operation (15 V) and low power consumption make the AD627 ideal for industrial applications, including 4 to 20 mA loop-powered systems. The AD627 does not compromise performance, unlike other micropower instrumentation amplifiers. Low voltage offset, offset drift, gain error, and gain drift minimize errors in the system. The AD627 also minimizes errors over frequency by providing excellent CMRR over frequency. Because the CMRR remains high up to 200 Hz, line noise and line harmonics are rejected. The AD627 provides superior performance, uses less circuit board area, and costs less than micropower discrete designs. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2007 Analog Devices, Inc. All rights reserved. AD627 TABLE OF CONTENTS Features .............................................................................................. 1 Applications....................................................................................... 1 Functional Block Diagram .............................................................. 1 General Description ......................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Single Supply ................................................................................. 3 Dual Supply ................................................................................... 5 Dual and Single Supplies ............................................................. 6 Absolute Maximum Ratings............................................................ 7 ESD Caution.................................................................................. 7 Pin Configurations and Function Descriptions ........................... 8 Typical Performance Characteristics ............................................. 9 Theory of Operation ...................................................................... 14 Using the AD627 ............................................................................ 15 Basic Connections ...................................................................... 15 Setting the Gain .......................................................................... 15 Reference Terminal .................................................................... 16 Input Range Limitations in Single-Supply Applications....... 16 Output Buffering ........................................................................ 17 Input and Output Offset Errors................................................ 17 Make vs. Buy: A Typical Application Error Budget............... 18 Errors Due to AC CMRR .......................................................... 19 Ground Returns for Input Bias Currents ................................ 19 Layout and Grounding .............................................................. 20 Input Protection ......................................................................... 21 RF Interference ........................................................................... 21 Applications Circuits...................................................................... 22 Classic Bridge Circuit ................................................................ 22 4 to 20 mA Single-Supply Receiver.......................................... 22 Thermocouple Amplifier .......................................................... 22 Outline Dimensions ....................................................................... 24 Ordering Guide .......................................................................... 24 REVISION HISTORY 11/07--Rev. C to Rev. D Changes to Features.......................................................................... 1 Changes to Figure 29 to Figure 34 Captions ............................... 13 Changes to Setting the Gain Section............................................ 15 Changes to Input Range Limitations in Single-Supply Applications Section....................................................................... 16 Changes to Table 7.......................................................................... 17 Changes to Figure 41...................................................................... 18 11/05--Rev. B to Rev. C Updated Format..................................................................Universal Added Pin Configurations and Function Descriptions Section ........................................................................ 8 Change to Figure 33 ....................................................................... 13 Updated Outline Dimensions ....................................................... 24 Changes to Ordering Guide .......................................................... 24 Rev. A to Rev. B Changes to Figure 4 and Table I, Resulting Gain column......... 11 Change to Figure 9 ......................................................................... 13 Rev. D | Page 2 of 24 AD627 SPECIFICATIONS SINGLE SUPPLY Typical @ 25C single supply, VS = 3 V and 5 V, and RL = 20 k, unless otherwise noted. Table 1. Parameter GAIN Gain Range Gain Error 1 G = +5 G = +10 G = +100 G = +1000 Nonlinearity G = +5 G = +100 Gain vs. Temperature1 G = +5 G > +5 VOLTAGE OFFSET Input Offset, VOSI 2 Over Temperature Average TC Output Offset, VOSO Over Temperature Average TC Offset Referred to the Input vs. Supply (PSRR) G = +5 G = +10 G = +100 G = +1000 INPUT CURRENT Input Bias Current Over Temperature Average TC Input Offset Current Over Temperature Average TC INPUT Input Impedance Differential Common-Mode Input Voltage Range 3 Common-Mode Rejection Ratio3 DC to 60 Hz with 1 k Source Imbalance G = +5 G = +5 OUTPUT Output Swing Short-Circuit Current Conditions G = +5 + (200 k/RG) VOUT = (-VS) + 0.1 to (+VS) - 0.15 0.03 0.15 0.15 0.50 10 20 10 -75 50 VCM = VREF = +VS/2 0.1 0.10 0.35 0.35 0.70 100 100 20 0.01 0.10 0.10 0.25 10 20 10 -75 25 0.1 0.06 0.25 0.25 0.35 100 100 20 % % % % ppm ppm ppm/C ppm/C V V V/C V V V/C Min 5 AD627A Typ Max 1000 Min 5 AD627B Typ Max 1000 Unit V/V 2.5 250 445 3 1000 1650 10 2.5 150 215 1 500 1150 10 86 100 110 110 100 120 125 125 3 20 0.3 1 10 15 1 2 86 100 110 110 100 120 125 125 3 20 0.3 1 10 15 1 2 dB dB dB dB nA nA pA/C nA nA pA/C 20||2 20||2 VS = 2.2 V to 36 V VREF = VS/2 (-VS) - 0.1 (+VS) - 1 (-VS) - 0.1 20||2 20||2 (+VS) - 1 G||pF G||pF V VS = 3 V, VCM = 0 V to 1.9 V VS = 5 V, VCM = 0 V to 3.7 V RL = 20 k RL = 100 k Short circuit to ground 77 77 (-VS) + 25 (-VS) + 7 90 90 (+VS) - 70 (+VS) - 25 25 83 83 (-VS) + 25 (-VS) + 7 96 96 (+VS) - 70 (+VS) - 25 25 dB dB mV mV mA Rev. D | Page 3 of 24 AD627 Parameter DYNAMIC RESPONSE Small Signal -3 dB Bandwidth G = +5 G = +100 G = +1000 Slew Rate Settling Time to 0.01% G = +5 G = +100 Settling Time to 0.01% G = +5 G = +100 Overload Recovery 1 2 Conditions Min AD627A Typ Max Min AD627B Typ Max Unit 80 3 0.4 +0.05/-0.07 VS = 3 V, 1.5 V output step 65 290 VS = 5 V, 2.5 V output step 85 330 3 80 3 0.4 +0.05/-0.07 65 290 85 330 3 kHz kHz kHz V/s s s s s s 50% input overload Does not include effects of External Resistor RG. See Table 8 for total RTI errors. 3 See the Using the AD627 section for more information on the input range, gain range, and common-mode range. Rev. D | Page 4 of 24 AD627 DUAL SUPPLY Typical @ 25C dual supply, VS = 5 V and 15 V, and RL = 20 k, unless otherwise noted. Table 2. Parameter GAIN Gain Range Gain Error 1 G = +5 G = +10 G = +100 G = +1000 Nonlinearity G = +5 G = +100 Gain vs. Temperature1 G = +5 G > +5 VOLTAGE OFFSET Input Offset, VOSI 2 Over Temperature Average TC Output Offset, VOSO Over Temperature Average TC Offset Referred to the Input vs. Supply (PSRR) G = +5 G = +10 G = +100 G = +1000 INPUT CURRENT Input Bias Current Over Temperature Average TC Input Offset Current Over Temperature Average TC INPUT Input Impedance Differential Common Mode Input Voltage Range 3 Common-Mode Rejection Ratio3 DC to 60 Hz with 1 k Source Imbalance G = +5 to +1000 G = +5 to +1000 OUTPUT Output Swing Short-Circuit Current Conditions G = +5 + (200 k/RG) VOUT = (-VS) + 0.1 to (+VS) - 0.15 0.03 0.15 0.15 0.50 VS = 5 V/15 V VS = 5 V/15 V 10/25 10/15 10 -75 Total RTI error = VOSI + VOSO/G 25 VCM = VREF = 0 V 0.1 200 395 3 1000 1700 10 25 0.1 125 190 1 500 1100 10 V V V/C V V V/C 0.10 0.35 0.35 0.70 100 100 20 0.01 0.10 0.10 0.25 10/25 10/15 10 -75 0.06 0.25 0.25 0.35 100 100 20 % % % % ppm ppm ppm/C ppm/C Min 5 AD627A Typ Max 1000 Min 5 AD627B Typ Max 1000 Unit V/V 2.5 2.5 86 100 110 110 100 120 125 125 2 20 0.3 5 10 15 1 5 86 100 110 110 100 120 125 125 2 20 0.3 5 10 15 1 5 dB dB dB dB nA nA pA/C nA nA pA/C 20||2 20||2 VS = 1.1 V to 18 V (-VS) - 0.1 (+VS) - 1 (-VS) - 0.1 20||2 20||2 (+VS) - 1 G||pF G||pF V VS = 5 V, VCM = -4 V to +3.0 V VS = 15 V, VCM = -12 V to +10.9 V RL = 20 k RL = 100 k Short circuit to ground 77 77 90 90 83 83 96 96 dB dB (-VS) + 25 (-VS) + 7 25 Rev. D | Page 5 of 24 (+VS) - 70 (+VS) - 25 (-VS) + 25 (-VS) + 7 25 (+VS) - 70 (+VS) - 25 mV mV mA AD627 Parameter DYNAMIC RESPONSE Small Signal -3 dB Bandwidth G = +5 G = +100 G = +1000 Slew Rate Settling Time to 0.01% G = +5 G = +100 Settling Time to 0.01% G = +5 G = +100 Overload Recovery 1 2 Conditions Min AD627A Typ Max Min AD627B Typ Max Unit 80 3 0.4 +0.05/-0.06 VS = 5 V, +5 V output step 135 350 VS = 15 V, +15 V output step 330 560 3 330 560 3 80 3 0.4 +0.05/-0.06 kHz kHz kHz V/s 135 350 s s 50% input overload s s s Does not include effects of External Resistor RG. See Table 8 for total RTI errors. 3 See the Using the AD627 section for more information on the input range, gain range, and common-mode range. DUAL AND SINGLE SUPPLIES Table 3. Parameter NOISE Voltage Noise, 1 kHz Input, Voltage Noise, eni Output, Voltage Noise, eno RTI, 0.1 Hz to 10 Hz G = +5 G = +1000 Current Noise 0.1 Hz to 10 Hz REFERENCE INPUT RIN Gain to Output Voltage Range 1 POWER SUPPLY Operating Range Quiescent Current Over Temperature TEMPERATURE RANGE For Specified Performance 1 Conditions Total RTI Noise = Min AD627A Typ Max Min AD627B Typ Max Unit (eni )2 + (eno / RG )2 38 177 1.2 0.56 50 1.0 125 1 38 177 1.2 0.56 50 1.0 125 1 nV/Hz nV/Hz V p-p V p-p fA/Hz pA p-p k f = 1 kHz RG = Dual supply Single supply 1.1 2.2 60 200 -40 18 36 85 1.1 2.2 60 200 -40 18 36 85 V V A nA/C C +85 +85 See Using the AD627 section for more information on the reference terminal, input range, gain range, and common-mode range. Rev. D | Page 6 of 24 AD627 ABSOLUTE MAXIMUM RATINGS Table 4. Parameter Supply Voltage Internal Power Dissipation 1 PDIP (N-8) SOIC_N (R-8) -IN, +IN Common-Mode Input Voltage Differential Input Voltage (+IN - (-IN)) Output Short-Circuit Duration Storage Temperature Range (N, R) Operating Temperature Range Lead Temperature (Soldering, 10 sec) 1 Rating 18 V 1.3 W 0.8 W -VS - 20 V to +VS + 20 V -VS - 20 V to +VS + 20 V +VS - (-VS) Indefinite -65C to +125C -40C to +85C 300C Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ESD CAUTION Specification is for device in free air: 8-lead PDIP package: JA = 90C/W. 8-lead SOIC_N package: JA = 155C/W. Rev. D | Page 7 of 24 AD627 PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS RG -IN +IN -VS 1 2 3 4 8 AD627 TOP VIEW (Not to Scale) RG +VS 00782-051 RG -IN +IN -VS 1 2 3 4 8 7 6 5 AD627 TOP VIEW (Not to Scale) RG +VS REF 00782-052 7 6 5 OUTPUT REF OUTPUT Figure 3. 8-Lead PDIP Pin Configuration Figure 4. 8-Lead SOIC_N Pin Configuration Table 5. Pin Function Descriptions Pin No. 1 2 3 4 5 6 7 8 Mnemonic RG -IN +IN -VS REF OUTPUT +VS RG Description External Gain Setting Resistor. Place gain setting resistor across RG pins to set the gain. Negative Input. Positive Input. Negative Voltage Supply Pin. Reference Pin. Drive with low impedance voltage source to level shift the output voltage. Output Voltage. Positive Supply Voltage. External Gain Setting Resistor. Place gain setting resistor across RG pins to set the gain. Rev. D | Page 8 of 24 AD627 TYPICAL PERFORMANCE CHARACTERISTICS At 25C, VS = 5 V, RL = 20 k, unless otherwise noted. 100 90 80 70 60 50 40 30 20 GAIN = +1000 10 00782-003 -5.5 -5.0 INPUT BIAS CURRENT (nA) NOISE (nV/ Hz, RTI) -4.5 -4.0 -3.5 -3.0 -2.5 VS = 15V -2.0 00782-006 00782-008 VS = +5V VS = 5V GAIN = +5 GAIN = +100 0 1 10 100 1k FREQUENCY (Hz) 10k 100k -1.5 -60 -40 -20 0 20 40 60 TEMPERATURE (C) 80 100 120 140 Figure 5. Voltage Noise Spectral Density vs. Frequency 100 90 Figure 8. Input Bias Current vs. Temperature 65.5 70 60 50 40 30 20 10 00782-004 POWER SUPPLY CURRENT (A) 80 64.5 CURRENT NOISE (fA/ Hz) 63.5 62.5 61.5 60.5 1 10 100 FREQUENCY (Hz) 1k 10k 0 5 10 15 20 25 30 TOTAL POWER SUPPLY VOLTAGE (V) 35 40 Figure 6. Current Noise Spectral Density vs. Frequency -3.2 Figure 9. Supply Current vs. Supply Voltage V+ VS = 15V (V+) -1 OUTPUT VOLTAGE SWING (V) -3.0 INPUT BIAS CURRENT (nA) -2.8 (V+) -2 VS = 1.5V VS = 2.5V SOURCING VS = 5V -2.6 (V+) -3 -2.4 (V-) +2 SINKING VS = 2.5V VS = 5V VS = 15V 10 15 OUTPUT CURRENT (mA) 20 25 -2.2 (V-) +1 VS = 1.5V -10 0 -5 5 COMMON-MODE INPUT (V) 10 15 00782-005 -2.0 -15 V- 0 5 Figure 7. Input Bias Current vs. CMV, VS = 15 V Figure 10. Output Voltage Swing vs. Output Current Rev. D | Page 9 of 24 00782-007 0 59.5 AD627 120 500mV 100 1s 110 100 POSITIVE PSRR (dB) G = +1000 G = +100 90 80 70 60 50 40 G = +5 10 00782-009 30 10 100 1k FREQUENCY (Hz) 10k 100k 00782-012 00782-014 20 Figure 11. 0.1 Hz to 10 Hz Current Noise (0.71 pA/DIV) 100 20mV 100 Figure 14. Positive PSRR vs. Frequency, 5 V 1s 1s 90 80 NEGATIVE PSRR (dB) 70 60 50 40 30 G = +5 20 G = +100 G = +1000 10 00782-010 10 10 100 1k FREQUENCY (Hz) 10k 100k 00782-013 0 Figure 12. 0.1 Hz to 10 Hz RTI Voltage Noise (400 nV/DIV), G = +5 120 2V 100 Figure 15. Negative PSRR vs. Frequency, 5 V 1s 110 100 G = +1000 POSITIVE PSRR (dB) 90 80 70 60 50 40 G = +100 G = +5 10 00782-011 30 20 10 100 1k FREQUENCY (Hz) 10k 100k Figure 13. 0.1 Hz to 10 Hz RTI Voltage Noise (200 nV/DIV), G = +1000 Figure 16. Positive PSRR vs. Frequency (VS = 5 V, 0 V) Rev. D | Page 10 of 24 AD627 10 400 300 SETTLING TIME (ms) 1 SETTLING TIME (s) 200 100 00782-015 5 10 100 GAIN (V/V) 1k 0 2 4 6 OUTPUT PULSE (V) 8 10 Figure 17. Settling Time to 0.01% vs. Gain for a 5 V Step at Output, RL = 20 k, CL = 100 pF, VS = 5 V 1mV 1V 50s Figure 20. Settling Time to 0.01% vs. Output Swing, G = +5, RL = 20 k, CL = 100 pF 200V 1V 100s Figure 18. Large Signal Pulse Response and Settling Time, G = -5, RL = 20 k, CL = 100 pF (1.5 mV = 0.01%) 1mV 1V 50s 00782-016 Figure 21. Large Signal Pulse Response and Settling Time, G = -100, RL = 20 k, CL = 100 pF (100 V = 0.01%) 200V 1V 500s Figure 19. Large Signal Pulse Response and Settling Time, G = -10, RL = 20 k, CL = 100 pF (1.0 mV = 0.01%) Figure 22. Large Signal Pulse Response and Settling Time, G = -1000, RL = 20 k, CL = 100 pF (10 V = 0.01%) Rev. D | Page 11 of 24 00782-020 00782-017 00782-019 00782-018 0.1 0 AD627 120 110 100 90 80 CMRR (dB) CH2 20mV A 20s 286mV EXT1 G = +1000 70 60 50 40 30 G = +100 G = +5 10 1 10 100 1k FREQUENCY (Hz) 10k 100k 00782-021 0 Figure 23. CMRR vs. Frequency, 5 VS (CMV = 200 mV p-p) 70 60 50 40 GAIN (dB) Figure 26. Small Signal Pulse Response, G = +10, RL = 20 k, CL = 50 pF G = +1000 CH2 20mV A 100s 286mV EXT1 G = +100 30 20 10 0 -10 -20 1k 10k FREQUENCY (Hz) 100k 00782-022 G = +10 G = +5 -30 100 Figure 24. Gain vs. Frequency (VS = 5 V, 0 V), VREF = 2.5 V Figure 27. Small Signal Pulse Response, G = +100, RL = 20 k, CL = 50 pF CH2 20mV A 20s 288mV EXT1 CH2 50mV A 1ms 286mV EXT1 00782-023 Figure 25. Small Signal Pulse Response, G = +5, RL = 20 k, CL = 50 pF Figure 28. Small Signal Pulse Response, G = +1000, RL = 20 k, CL = 50 pF Rev. D | Page 12 of 24 00782-026 00782-025 00782-024 20 AD627 20V/DIV 200V/DIV 00782-027 Figure 29. Gain Nonlinearity, Negative Input, VS = 2.5 V, G = +5 (4 ppm/DIV) 40V/DIV Figure 32. Gain Nonlinearity, Negative Input, VS = 15 V, G = +100 (7 ppm/DIV) 200V/DIV 00782-028 Figure 30. Gain Nonlinearity, Negative Input, VS = 2.5 V, G = +100 (8 ppm/DIV) 40V/DIV Figure 33. Gain Nonlinearity, Negative Input, VS = 15 V, G = +5 (7 ppm/DIV) 200V/DIV 00782-029 Figure 31. Gain Nonlinearity, Negative Input, VS = 15 V, G = +5 (1.5 ppm/DIV) Figure 34. Gain Nonlinearity, Negative Input, VS = 15 V, G = +100 (7 ppm/DIV) Rev. D | Page 13 of 24 00782-032 VOUT 3V/DIV VOUT 3V/DIV 00782-031 VOUT 0.5V/DIV VOUT 3V/DIV 00782-030 VOUT 0.5V/DIV VOUT 3V/DIV AD627 THEORY OF OPERATION The AD627 is a true instrumentation amplifier, built using two feedback loops. Its general properties are similar to those of the classic two-op-amp instrumentation amplifier configuration but internally the details are somewhat different. The AD627 uses a modified current feedback scheme, which, coupled with interstage feedforward frequency compensation, results in a much better common-mode rejection ratio (CMRR) at frequencies above dc (notably the line frequency of 50 Hz to 60 Hz) than might otherwise be expected of a low power instrumentation amplifier. In Figure 35, A1 completes a feedback loop that, in conjunction with V1 and R5, forces a constant collector current in Q1. Assume that the gain-setting resistor (RG) is not present. Resistors R2 and R1 complete the loop and force the output of A1 to be equal to the voltage on the inverting terminal with a gain of nearly 1.25. A2 completes a nearly identical feedback loop that forces a current in Q2 that is nearly identical to that in Q1; A2 also provides the output voltage. When both loops are balanced, the gain from the noninverting terminal to VOUT is equal to 5, whereas the gain from the output of A1 to VOUT is equal to -4. The inverting terminal gain of A1 (1.25) times the gain of A2 (-4) makes the gain from the inverting and noninverting terminals equal. The differential mode gain is equal to 1 + R4/R3, nominally 5, and is factory trimmed to 0.01% final accuracy. Adding an external gain setting resistor (RG) increases the gain by an amount equal to (R4 + R1)/RG. The output voltage of the AD627 is given by VOUT = [VIN(+) - VIN(-)] x (5 + 200 k/RG) + VREF (1) Laser trims are performed on R1 through R4 to ensure that their values are as close as possible to the absolute values in the gain equation. This ensures low gain error and high commonmode rejection at all practical gains. EXTERNAL GAIN RESISTOR R1 100k +VS -IN 2k Q1 R2 25k RG R3 25k R4 100k +VS Q2 2k +IN REF -VS A1 -VS A2 OUTPUT 00782-033 R5 200k V1 0.1V R6 200k Figure 35. Simplified Schematic Rev. D | Page 14 of 24 AD627 USING THE AD627 BASIC CONNECTIONS Figure 36 shows the basic connection circuit for the AD627. The +VS and -VS terminals connect to the power supply. The supply can be either bipolar (VS = 1.1 V to 18 V) or single supply (-VS = 0 V, +VS = 2.2 V to 36 V). Capacitively decouple the power supplies close to the power pins of the device. For best results, use surface-mount 0.1 F ceramic chip capacitors. The input voltage can be single-ended (tie either -IN or +IN to ground) or differential. The difference between the voltage on the inverting and noninverting pins is amplified by the programmed gain. The gain resistor programs the gain as described in the Setting the Gain and Reference Terminal sections. Basic connections are shown in Figure 36. The output signal appears as the voltage difference between the output pin and the externally applied voltage on the REF pin, as shown in Figure 37. SETTING THE GAIN The gain of the AD627 is resistor programmed by RG, or, more precisely, by whatever impedance appears between Pin 1 and Pin 8. The gain is set according to Gain = 5 + (200 k/RG) or RG = 200 k/(Gain - 5) (2) Therefore, the minimum achievable gain is 5 (for 200 k/ (Gain - 5)). With an internal gain accuracy of between 0.05% and 0.7%, depending on gain and grade, a 0.1% external gain resistor is appropriate to prevent significant degradation of the overall gain error. However, 0.1% resistors are not available in a wide range of values and are quite expensive. Table 6 shows recommended gain resistor values using 1% resistors. For all gains, the size of the gain resistor is conservatively chosen as the closest value from the standard resistor table that is higher than the ideal value. This results in a gain that is always slightly less than the desired gain, thereby preventing clipping of the signal at the output due to resistor tolerance. The internal resistors on the AD627 have a negative temperature coefficient of -75 ppm/C maximum for gains > 5. Using a gain resistor that also has a negative temperature coefficient of -75 ppm/C or less tends to reduce the overall gain drift of the circuit. +VS +1.1V TO +18V 0.1F +IN +VS +2.2V TO +36V 0.1F +IN VIN RG -IN RG OUTPUT RG REF 0.1F VOUT REF (INPUT) VIN RG -IN RG OUTPUT RG REF VOUT REF (INPUT) -VS -1.1V TO -18V GAIN = 5 + (200k/RG) Figure 36. Basic Connections for Single and Dual Supplies V+ VDIFF 2 +IN 100k EXTERNAL GAIN RESISTOR 25k RG 25k +VS Q1 -VS Q2 -VS A2 200k 0.1V VA 200k -VS OUTPUT 00782-035 REF VCM VDIFF 2 100k +VS -IN V- A1 -IN 2k 2k +IN Figure 37. Amplifying Differential Signals with a Common-Mode Component Rev. D | Page 15 of 24 00782-034 AD627 Table 6. Recommended Values of Gain Resistors Desired Gain 5 6 7 8 9 10 15 20 25 30 40 50 60 70 80 90 100 200 500 1000 1% Standard Table Value of RG 200 k 100 k 68.1 k 51.1 k 40.2 k 20 k 13.7 k 10 k 8.06 k 5.76 k 4.53 k 3.65 k 3.09 k 2.67 k 2.37 k 2.1 k 1.05 k 412 205 Resulting Gain 5.00 6.00 7.00 7.94 8.91 9.98 15.00 19.60 25.00 29.81 39.72 49.15 59.79 69.72 79.91 89.39 100.24 195.48 490.44 980.61 The voltage on A1 can also be expressed as a function of the actual voltages on the -IN and +IN pins (V- and V+) such that VA1 = 1.25 ((V-) + 0.5 V) - 0.25 VREF - ((V+) - (V-)) 25 k/RG (4) The output of A1 is capable of swinging to within 50 mV of the negative rail and to within 200 mV of the positive rail. It is clear, from either Equation 3 or Equation 4, that an increasing VREF (while it acts as a positive offset at the output of the AD627) tends to decrease the voltage on A1. Figure 38 and Figure 39 show the maximum voltages that can be applied to the REF pin for a gain of 5 for both the single-supply and dual-supply cases. 5 4 3 2 MAXIMUM VREF VREF (V) 1 0 -1 -2 -3 -4 -5 -4 -3 -2 0 -1 VIN(-) (V) 1 2 3 4 00782-036 00782-037 MINIMUM VREF -5 -6 REFERENCE TERMINAL The reference terminal potential defines the zero output voltage and is especially useful when the load does not share a precise ground with the rest of the system. It provides a direct means of injecting a precise offset to the output. The reference terminal is also useful when amplifying bipolar signals, because it provides a virtual ground voltage. The AD627 output voltage is developed with respect to the potential on the reference terminal; therefore, tying the REF pin to the appropriate local ground solves many grounding problems. For optimal CMR, tie the REF pin to a low impedance point. Figure 38. Reference Input Voltage vs. Negative Input Voltage, VS = 5 V, G = +5 5 4 MAXIMUM VREF VREF (V) 3 2 MINIMUM VREF 1 INPUT RANGE LIMITATIONS IN SINGLE-SUPPLY APPLICATIONS In general, the maximum achievable gain is determined by the available output signal range. However, in single-supply applications where the input common-mode voltage is nearly or equal to 0, some limitations on the gain can be set. Although the Specifications section nominally defines the input, output, and reference pin ranges, the voltage ranges on these pins are mutually interdependent. Figure 37 shows the simplified schematic of the AD627, driven by a differential voltage (VDIFF) that has a common-mode component, VCM. The voltage on the A1 op amp output is a function of VDIFF, VCM, the voltage on the REF pin, and the programmed gain. This voltage is given by VA1 = 1.25 (VCM + 0.5 V) - 0.25 VREF - VDIFF (25 k/RG - 0.625) (3) 0 -0.5 0 0.5 1.0 1.5 2.0 2.5 VIN(-) (V) 3.0 3.5 4.0 4.5 Figure 39. Reference Input Voltage vs. Negative Input Voltage, VS = 5 V, G = +5 Raising the input common-mode voltage increases the voltage on the output of A1. However, in single-supply applications where the common-mode voltage is low, a differential input voltage or a voltage on REF that is too high can drive the output of A1 into the ground rail. Some low-side headroom is added because both inputs are shifted upwards by about 0.5 V (that is, by the VBE of Q1 and Q2). Use Equation 3 and Equation 4 to check whether the voltage on Amplifier A1 is within its operating range. Rev. D | Page 16 of 24 AD627 Table 7. Maximum Gain for Low Common-Mode, Single-Supply Applications VIN 100 mV, VCM = 0 V 50 mV, VCM = 0 V 10 mV, VCM = 0 V V- = 0 V, V+ = 0 V to 1 V V- = 0 V, V+ = 0 mV to 100 mV V- = 0 V, V+ = 0 mV to 10 mV REF Pin 2V 2V 2V 1V 1V 1V Supply Voltage 5 V to 15 V 5 V to 15 V 5 V to 15 V 10 V to 15 V 5 V to 15 V 5 V to 15 V RG (1% Tolerance) 28.7 k 10.7 k 1.74 k 78.7 k 7.87 k 787 Resulting Maximum Gain 12.0 23.7 119.9 7.5 31 259.1 Output Swing WRT 0 V 0.8 V to 3.2 V 0.8 V to 3.2 V 0.8 V to 3.2 V 1 V to 8.5 V 1 V to 4.1 V 1 V to 3.6 V Table 8. RTI Error Sources Gain +5 +10 +20 +50 +100 +500 +1000 Maximum Total RTI Offset Error (V) AD627A AD627B 450 250 350 200 300 175 270 160 270 155 252 151 251 151 Maximum Total RTI Offset Drift (V/C) AD627A AD627B 5 3 4 2 3.5 1.5 3.2 1.2 3.1 1.1 3 1 3 1 Total RTI Noise (nV/Hz) AD627A /AD627B 95 66 56 53 52 52 52 Table 7 gives values for the maximum gain for various singlesupply input conditions. The resulting output swings refer to 0 V. To maximize the available gain and output swing, set the voltages on the REF pins to either 2 V or 1 V. In most cases, there is no advantage to increasing the single supply to greater than 5 V (the exception is an input range of 0 V to 1 V). INPUT AND OUTPUT OFFSET ERRORS The low errors of the AD627 are attributed to two sources, input and output errors. The output error is divided by G when referred to the input. In practice, input errors dominate at high gains and output errors dominate at low gains. The total offset error for a given gain is calculated as Total Error RTI = Input Error + (Output Error/Gain) Total Error RTO = (Input Error x G) + Output Error (5) (6) OUTPUT BUFFERING The AD627 is designed to drive loads of 20 k or greater but can deliver up to 20 mA to heavier loads at lower output voltage swings (see Figure 10). If more than 20 mA of output current is required at the output, buffer the AD627 output with a precision op amp, such as the OP113. Figure 40 shows this for a single supply. This op amp can swing from 0 V to 4 V on its output while driving a load as small as 600 . +VS 0.1F 0.1F VIN RG RTI offset errors and noise voltages for different gains are listed in Table 8. AD627 REF 0.1F RG OP113 0.1F VOUT -VS -VS 00782-038 Figure 40. Output Buffering Rev. D | Page 17 of 24 AD627 MAKE vs. BUY: A TYPICAL APPLICATION ERROR BUDGET The example in Figure 41 serves as a good comparison between the errors associated with an integrated and a discrete in-amp implementation. A 100 mV signal from a resistive bridge (common-mode voltage = 2.5 V) is amplified. This example compares the resulting errors from a discrete two-op-amp instrumentation amplifier and the AD627. The discrete implementation uses a four-resistor precision network (1% match, 50 ppm/C tracking). The errors associated with each implementation (see Table 9) show the integrated in-amp to be more precise at both ambient and overtemperature. Note that the discrete implementation is more expensive, primarily due to the relatively high cost of the low drift precision resistor network. The input offset current of the discrete instrumentation amplifier implementation is the difference in the bias currents of the twoop amplifiers, not the offset currents of the individual op amps. In addition, although the values of the resistor network are chosen so that the inverting and noninverting inputs of each op amp see the same impedance (about 350 ), the offset current of each op amp adds another error that must be characterized. +5V 350 350 +5V +5V LT10781SB 350 350 100mV RG 40.2k 1% +10ppm/C AD627A 1/2 VOUT 1/2 +2.5V LT10781SB VOUT AD627A GAIN = 9.98 (5+(200k/R G)) HOMEBREW IN-AMP, G = +10 *1% RESISTOR MATCH, 50ppm/C TRACKING Figure 41. Make vs. Buy Table 9. Make vs. Buy Error Budget Total Error AD627 (ppm) 3,500 3.5 Total Error Homebrew (ppm) 3,600 70 2.45 25,000 10,000 38,672 3,000 4,200 7 7,207 45,879 Error Source ABSOLUTE ACCURACY at TA = 25C Total RTI Offset Voltage, mV Input Offset Current, nA Internal Offset Current (Homebrew Only) CMRR, dB Gain DRIFT TO 85C Gain Drift, ppm/C Total RTI Offset Voltage, mV/C Input Offset Current, pA/C AD627 Circuit Calculation (250 V + (1000 V/10))/100 mV 1 nA x 350 /100 mV Not applicable 77 dB141 ppm x 2.5 V/100 mV 0.35% + 0.1% Homebrew Circuit Calculation (180 V x 2)/100 mV 20 nA x 350 /100 mV 0.7 nA x 350 /100 mV (1% match x 2.5 V)/10/100 mV 1% match Total Absolute Error 50 ppm/C x 60C (2 x 3.5 V/C x 60C)/100 mV (33 pA/C x 350 x 60C)/100 mV Total Drift Error Grand Total Error 3,531 13,500 20,535 3,900 2,600 3.5 6,504 27,039 (-75 + 10) ppm/C x 60C (3.0 V/C + (10 V/C/10)) x 60C/100 mV (16 pA/C x 350 x 60C)/100 mV Rev. D | Page 18 of 24 00782-039 +2.5V 3.15k* 350* 350* 3.15k* AD627 ERRORS DUE TO AC CMRR In Table 9, the error due to common-mode rejection results from the common-mode voltage from the bridge 2.5 V. The ac error due to less than ideal common-mode rejection cannot be calculated without knowing the size of the ac common-mode voltage (usually interference from 50 Hz/60 Hz mains frequencies). A mismatch of 0.1% between the four gain setting resistors determines the low frequency CMRR of a two-op-amp instrumentation amplifier. The plot in Figure 43 shows the practical results of resistor mismatch at ambient temperature. The CMRR of the circuit in Figure 42 (Gain = +11) was measured using four resistors with a mismatch of nearly 0.1% (R1 = 9999.5 , R2 = 999.76 , R3 = 1000.2 , R4 = 9997.7 ). As expected, the CMRR at dc was measured at about 84 dB (calculated value is 85 dB). However, as frequency increases, CMRR quickly degrades. For example, a 200 mV p-p harmonic of the mains frequency at 180 Hz would result in an output voltage of about 800 V. To put this in context, a 12-bit data acquisition system, with an input range of 0 V to 2.5 V, has an LSB weighting of 610 V. By contrast, the AD627 uses precision laser trimming of internal resistors, along with patented CMR trimming, to yield a higher dc CMRR and a wider bandwidth over which the CMRR is flat (see Figure 23). +5V GROUND RETURNS FOR INPUT BIAS CURRENTS Input bias currents are dc currents that must flow to bias the input transistors of an amplifier. They are usually transistor base currents. When amplifying floating input sources, such as transformers or ac-coupled sources, there must be a direct dc path into each input so that the bias current can flow. Figure 44, Figure 45, and Figure 46 show how to provide a bias current path for the cases of, respectively, transformer coupling, a thermocouple application, and capacitive ac-coupling. In dc-coupled resistive bridge applications, providing this path is generally not necessary because the bias current simply flows from the bridge supply through the bridge and into the amplifier. However, if the impedance that the two inputs see are large, and differ by a large amount (>10 k), the offset current of the input stage causes dc errors compatible with the input offset voltage of the amplifier. -INPUT +VS 2 1 7 RG 8 AD627 5 4 3 6 VOUT +INPUT REFERENCE LOAD 00782-042 -VS TO POWER SUPPLY GROUND Figure 44. Ground Returns for Bias Currents with Transformer Coupled Inputs -INPUT +VS 2 1 7 VIN- VIN+ A1 1/2 OP296 A2 1/2 OP296 VOUT RG 8 AD627 5 4 3 6 VOUT +INPUT REFERENCE LOAD 00782-043 00782-044 00782-040 R1 9999.5 -5V R2 999.76 -VS R3 1000.2 R4 9997.7 TO POWER SUPPLY GROUND Figure 45. Ground Returns for Bias Currents with Thermocouple Inputs -INPUT +VS 2 1 7 Figure 42. 0.1% Resistor Mismatch Example 120 110 100 CMRR (dB) RG +INPUT 100k 8 3 AD627 5 4 6 VOUT REFERENCE LOAD 90 80 70 60 50 40 30 1 10 100 1k FREQUENCY (Hz) 10k 100k 00782-041 -VS TO POWER SUPPLY GROUND Figure 46. Ground Returns for Bias Currents with AC-Coupled Inputs 20 Figure 43. CMRR over Frequency of Discrete In-Amp in Figure 42 Rev. D | Page 19 of 24 AD627 LAYOUT AND GROUNDING The use of ground planes is recommended to minimize the impedance of ground returns (and hence, the size of dc errors). To isolate low level analog signals from a noisy digital environment, many data acquisition components have separate analog and digital ground returns (see Figure 47). Return all ground pins from mixed-signal components, such as analog-to-digital converters, through the high quality analog ground plane. Digital ground lines of mixed-signal components should also be returned through the analog ground plane. This may seem to break the rule of separating analog and digital grounds; however, in general, there is also a requirement to keep the voltage difference between digital and analog grounds on a converter as small as possible (typically, <0.3 V). The increased noise, caused by the digital return currents of the converter flowing through the analog ground plane, is generally negligible. To maximize isolation between analog and digital, connect the ground planes back at the supplies. If there is only one power supply available, it must be shared by both digital and analog circuitry. Figure 48 shows how to minimize interference between the digital and analog circuitry. As in the previous case, use separate analog and digital ground planes or use reasonably thick traces as an alternative to a digital ground plane. Connect the ground planes at the ground pin of the power supply. Run separate traces (or power planes) from the power supply to the supply pins of the digital and analog circuits. Ideally, each device should have its own power supply trace, but they can be shared by multiple devices if a single trace is not used to route current to both digital and analog circuitry. ANALOG POWER SUPPLY +5V -5V GND DIGITAL POWER SUPPLY GND +5V 0.1F 0.1F 0.1F 0.1F 7 2 4 1 6 4 3 6 14 AD627 3 5 VIN1 VIN2 VDD ADC AGND DGND 12 AGND VDD 00782-045 AD7892-2 MICROPROCESSOR Figure 47. Optimal Grounding Practice for a Bipolar Supply Environment with Separate Analog and Digital Supplies POWER SUPPLY 5V GND 0.1F 0.1F 0.1F 7 2 4 1 6 4 AD627 3 5 VIN VDD ADC AGND DGND 12 VDD DGND 00782-046 AD7892-2 MICROPROCESSOR Figure 48. Optimal Ground Practice in a Single-Supply Environment Rev. D | Page 20 of 24 AD627 INPUT PROTECTION As shown in the simplified schematic (see Figure 35), both the inverting and noninverting inputs are clamped to the positive and negative supplies by ESD diodes. In addition, a 2 k series resistor on each input provides current limiting in the event of an overvoltage. These ESD diodes can tolerate a maximum continuous current of 10 mA. So an overvoltage (that is, the amount by which the input voltage exceeds the supply voltage) of 20 V can be tolerated. This is true for all gains, and for power on and off. This last case is particularly important because the signal source and amplifier can be powered separately. If the overvoltage is expected to exceed 20 V, use additional external series current-limiting resistors to keep the diode current below 10 mA. Capacitor C3 is needed to maintain common-mode rejection at low frequencies. R1/R2 and C1/C2 form a bridge circuit whose output appears across the input pins of the in-amp. Any mismatch between C1 and C2 unbalances the bridge and reduces commonmode rejection. C3 ensures that any RF signals are common mode (the same on both in-amp inputs) and are not applied differentially. This second low-pass network, R1 + R2 and C3, has a -3 dB frequency equal to 1/(2((R1 + R2) x C3)) +VS 0.33F C1 R1 1000pF 20k 5% 1% +IN R2 C3 20k 0.022F 1% C2 1000pF 5% RG 0.01F (8) AD627 REFERENCE VOUT RF INTERFERENCE All instrumentation amplifiers can rectify high frequency outof-band signals. Once rectified, these signals appear as dc offset errors at the output. The circuit in Figure 49 provides good RFI suppression without reducing performance within the pass band of the instrumentation amplifier. Resistor R1 and Capacitor C1 (and likewise, R2 and C2) form a low-pass RC filter that has a -3 dB BW equal to f = 1/(2(R1 x C1)) (7) Using the component values shown in Figure 49, this filter has a -3 dB bandwidth of approximately 8 kHz. Resistor R1 and Resistor R2 were selected to be large enough to isolate the circuit input from the capacitors but not large enough to significantly increase circuit noise. To preserve common-mode rejection in the amplifier pass band, Capacitor C1 and Capacitor C2 must be 5% mica units, or low cost 20% units can be tested and binned to provide closely matched devices. -IN 0.33F -VS 0.01F 00782-047 Figure 49. Circuit to Attenuate RF Interference Using a C3 value of 0.022 F, as shown in Figure 49, the -3 dB signal bandwidth of this circuit is approximately 200 Hz. The typical dc offset shift over frequency is less than 1 mV and the RF signal rejection of the circuit is better than 57 dB. To increase the 3 dB signal bandwidth of this circuit, reduce the value of Resistor R1 and Resistor R2. The performance is similar to that when using 20 k resistors, except that the circuitry preceding the in-amp must drive a lower impedance load. When building a circuit like that shown in Figure 49, use a PC board with a ground plane on both sides. Make all component leads as short as possible. Resistor R1 and Resistor R2 can be common 1% metal film units, but Capacitor C1 and Capacitor C2 must be 5% tolerance devices to avoid degrading the commonmode rejection of the circuit. Either the traditional 5% silver mica units or Panasonic 2% PPS film capacitors are recommended. Rev. D | Page 21 of 24 AD627 APPLICATIONS CIRCUITS CLASSIC BRIDGE CIRCUIT Figure 50 shows the AD627 configured to amplify the signal from a classic resistive bridge. This circuit works in dual-supply mode or single-supply mode. Typically, the same voltage that powers the instrumentation amplifiers excites the bridge. Connecting the bottom of the bridge to the negative supply of the instrumentation amplifiers (usually 0 V, -5 V, -12 V, or -15 V), sets up an input common-mode voltage that is optimally located midway between the supply voltages. It is also appropriate to set the voltage on the REF pin to midway between the supplies, especially if the input signal is bipolar. However, the voltage on the REF pin can be varied to suit the application. For example, the REF pin is tied to the VREF pin of an analog-to-digital converter (ADC) whose input range is (VREF VIN). With an available output swing on the AD627 of (-VS + 100 mV) to (+VS - 150 mV), the maximum programmable gain is simply this output range divided by the input range. +VS 0.1F 4 TO 20 mA SINGLE-SUPPLY RECEIVER Figure 51 shows how a signal from a 4 to 20 mA transducer can be interfaced to the ADuC812, a 12-bit ADC with an embedded microcontroller. The signal from a 4 to 20 mA transducer is single-ended, which initially suggests the need for a simple shunt resistor to convert the current to a voltage at the high impedance analog input of the converter. However, any line resistance in the return path (to the transducer) adds a current dependent offset error; therefore, the current must be sensed differentially. In this example, a 24.9 shunt resistor generates a maximum differential input voltage to the AD627 of between 100 mV (for 4 mA in) and 500 mV (for 20 mA in). With no gain resistor present, the AD627 amplifies the 500 mV input voltage by a factor of 5, to 2.5 V, the full-scale input voltage of the ADC. The zero current of 4 mA corresponds to a code of 819 and the LSB size is 610 A. THERMOCOUPLE AMPLIFIER Because the common-mode input range of the AD627 extends 0.1 V below ground, it is possible to measure small differential signals that have a low, or no, common-mode component. Figure 51 shows a thermocouple application where one side of the J-type thermocouple is grounded. Over a temperature range from -200C to +200C, the J-type thermocouple delivers a voltage ranging from -7.890 mV to +10.777 mV. A programmed gain on the AD627 of 100 (RG = 2.1 k) and a voltage on the AD627 REF pin of 2 V result in the output voltage of the AD627 ranging from 1.110 V to 3.077 V relative to ground. For a different input range or different voltage on the REF pin, it is important to verify that the voltage on Internal Node A1 (see Figure 37) is not driven below ground. This can be checked using the equations in the Input Range Limitations in Single-Supply Applications section. 5V 0.1F 00782-048 VDIFF RG = 200k GAIN-5 AD627 0.1F VOUT VREF -VS Figure 50. Classic Bridge Circuit J-TYPE THERMOCOUPLE RG 2.1k AD627 REF VOUT VREF 00782-050 Figure 51. Amplifying Bipolar Signals with Low Common-Mode Voltage Rev. D | Page 22 of 24 AD627 5V 5V 0.1F VREF 4-20mA TRANSDUCER LINE IMPEDANCE AVDD DVDD 0.1F 5V 0.1F 4-20mA 24.9 G = +5 AD627 REF AIN 0 to AIN 7 ADuC812 MICROCONVERTER(R) AGND DGND 00782-049 Figure 52. 4 to 20 mA Receiver Circuit Rev. D | Page 23 of 24 AD627 OUTLINE DIMENSIONS 0.400 (10.16) 0.365 (9.27) 0.355 (9.02) 8 1 5 4 0.280 (7.11) 0.250 (6.35) 0.240 (6.10) 0.100 (2.54) BSC 0.210 (5.33) MAX 0.150 (3.81) 0.130 (3.30) 0.115 (2.92) 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 0.070 (1.78) 0.060 (1.52) 0.045 (1.14) 0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.060 (1.52) MAX 0.195 (4.95) 0.130 (3.30) 0.115 (2.92) 5.00 (0.1968) 4.80 (0.1890) 4.00 (0.1574) 3.80 (0.1497) 8 1 5 4 6.20 (0.2441) 5.80 (0.2284) 0.015 (0.38) MIN SEATING PLANE 0.005 (0.13) MIN 0.015 (0.38) GAUGE PLANE 0.430 (10.92) MAX 0.014 (0.36) 0.010 (0.25) 0.008 (0.20) 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY 0.10 SEATING PLANE 1.75 (0.0688) 1.35 (0.0532) 0.50 (0.0196) 0.25 (0.0099) 8 0 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) 45 0.51 (0.0201) 0.31 (0.0122) COMPLIANT TO JEDEC STANDARDS MS-001 CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS. COMPLIANT TO JEDEC STANDARDS MS-012-A A CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. Figure 53. 8-Lead Plastic Dual In-Line Package [PDIP] Narrow Body (N-8) Dimensions shown in inches (and millimeters) Figure 54. 8-Lead Small Standard Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters (and inches) ORDERING GUIDE Model AD627AN AD627ANZ 1 AD627AR AD627AR-REEL AD627AR-REEL7 AD627ARZ1 AD627ARZ-R71 AD627ARZ-RL1 AD627BN AD627BNZ1 AD627BR AD627BR-REEL AD627BR-REEL7 AD627BRZ1 AD627BRZ-RL1 AD627BRZ-R71 1 Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C Package Description 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] 8-Lead Small Standard Outline [SOIC_N] Package Option N-8 N-8 R-8 R-8 R-8 R-8 R-8 R-8 N-8 N-8 R-8 R-8 R-8 R-8 R-8 R-8 Z = RoHS Compliant part. (c)2007 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D00782-0-11/07(D) Rev. D | Page 24 of 24 012407-A |
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